1. Field of the Invention
This invention relates to a spread spectrum demodulator of a direct sequence type spread spectrum system.
2. Description of Related Arts
A spread spectrum demodulator for use with a phase-shift-keying modulated spread spectrum signal disclosed, for example in the article entitled "YOKOYAMA SPREAD SPECTRUM COMMUNICATION SYSTEM" by Kagaku Gijyutsu Syuppan Sha (in 1988), has been known as a spread spectrum demodulator of the conventional direct sequence type spread spectrum system. FIG. 16 of the accompanying drawings diagrammatically represents a block diagram of the conventional spread spectrum demodulator.
In the drawing, reference numeral 1 designates a received signal; 2, an initial acquisition circuit for generating from the received signal a signal (hereinafter referred to as acquired signal) which represents a breakpoint of information data bits; 3, an acquired signal; 4, a tracking circuit for effecting tracking in response to the acquired signal; 5, a reproduced pseudo noise signal (hereinafter referred to as PN signal); 6, a mixer for multiplying the received signal by the reproduced PN signal; 7, a despreading section comprised of the initial acquisition circuit 2, the tracking circuit 4, and the mixer 6; 8, a despread signal which does not contain a PN signal component; 9, a PSK demodulator; and 10, demodulated data.
The outline of the operation of the demodulator shown in FIG. 16 will now be described. Assuming that the received signal is a spread spectrum signal whose carrier wave, having a frequency of .omega..sub.0, is modulated by BPSK (Binary Phase-shift Keying), the spread spectrum signal is then expressed as EQU d(t)c(t)cos.omega..sub.0 t (1).
In this equation, d(t) is an information data signal having a bit width of Td and is designated as a rectangular wave signal having a value of -1 or +1, and c(t) is a PN signal (hereinafter, one bit of the PN signal is called a chip) having the bit width of Tc and is designated as a rectangular wave signal having a value of -1 or +1. N chips of the PN signals constite one cycle of a M series code. Spreading one bit of information data with one cycle of the PN signal results in Td=NTc.
The initial acquisition circuit 2 detects from the received signal 1 a timing signal corresponding to a breakpoint of the information data bits and outputs the thus detected timing signal to the tracking circuit 4 as the acquired signal 3. The tracking circuit 4 generates the reproduced PN signal 5 from the acquired signal 3. Despite the fact that the reproduced PN signal 5 is out of phase with the PN signal included in the received signal 1 before a time when the acquired signal 3 is received, namely, c (t+.tau.), the tracking circuit 4 generates the reproduced PN signal 5 in synchronism with the received signal of .tau.=0 based on the timing when the acquired signal 3 is received, and thereafter tracks the status of .tau.=0.
The multiplier 6 multiplies the received signal 1 by the reproduced PN signal 5, which results in c(t).times.c(t)=1. Hence, there will be obtained the despread signal 8 which does not contain the PN signal component the despread signal being expressed as EQU d(t) cos .omega..sub.0 t (2)
The general PSK demodulator 9 demodulates the despread signal 8 and produces the demodulated data 10.
The feature of the spread spectrum demodulator lies in the despreading section 7 comprised of the initial acquisition circuit 2, the tracking circuit 4, and the mixer 6. The initial acquisition circuit 2 and the tracking circuit 4 both of which are components of the despreading section 7 will now be described with reference to FIGS. 17, 18, and 19.
FIG. 17 shows the structure of the initial acquisition circuit of the conventional spread spectrum demodulator disclosed in the book entitled "SPREAD SPECTRUM COMMUNICATIONS, Vol. 3", by M. K. Simon, COMPUTER SCIENCE PRESS (1985).
In the drawing, reference numeral 1 designates the received signal; 11, a carrier wave generator; 12A and 12B, mixers; 13A and 13B, low pass filters (LPF); 13a, a base band signal generated using the carrier wave which was generated by the carrier wave generator 11; 13b, a base band signal generated by use of the Carrier wave having a phase delay of 90.degree.; 14A and 14B, sampling circuits for sampling the base band signals 13a and 13b, respectively; 14a and 14b; sampled signals, each sampled by the sampling circuit; 15A and 15B, correlators for correlating the sampled signals 14a and 14b with a PN signal (hereinafter, referred to as reference PN signal) which has been preset by the demodulator; 15a and 15b, correlation signals outputted from the correlators; 15A and 15B respectively; 16A and 16B, square generators for producing a square of each of the correlation signals 15a and 15b; 17, a phase shifter for causing a 90.degree. phase delay of the carrier wave outputted from the carrier wave generator 11; 23, an adder for adding the squares of the correlation signals produced by the square generators 16A and 16B; 24, a correlation pulse signal outputted from the adder 23; and 25, a decision device for generating the acquired signal 3 by deciding on phase matching of the PN signal components included in the received signal 1 and the reference PN signal.
The operation of the initial acquisition circuit shown in FIG. 17 will now be described.
The function of the initial acquisition circuit is to generate the acquired signal 3 at a timing such that the PN signal included in the received signal 1 that was received with distortions due to adverse noise effects of along the transmission path is in phase with the reference PN signal. This is achieved by comparing these two signals.
Given that a phase difference between a carrier wave of the received signal 1 and a carrier wave outputted from the carrier wave generator 11 is 8, the carrier wave outputted from the carrier wave generator can be expressed as EQU cos(.omega..sub.0 t+.theta.).
In the meantime, a carrier wave outputted by way of the 90.degree. phase shifter 17 is EQU sin(.omega..sub.0 t+.theta.).
The received signal 1 is divided into two, and the one signal is multiplied by the carrier wave cos (.omega..sub.0 t+.theta.) which is outputted from the carrier wave generator 11 by means of the mixer 12A, then converted by means of the low pass filter 13A into the base band signal 13a which includes cos.theta. components. The other signal is multiplied by the carrier wave sin (.omega..sub.0 t+.theta.) that is outputted by way of the 90.degree. phase shifter 17 by means of the mixer 12B, then converted by means of the low pass filter 13B into the base band signal 13b which includes sin.theta. components.
The base band signals 13a and 13b are sampled by the sampling circuits 14A and 14B, and transmitted to the correlators 15A and 15B, respectively.
Because of the use of an asynchronous clock in the above sampling operation, sampling is usually performed twice per chip. Consequently, supposing a chip width Tc of the PN signal, a sampling interval will be given as Tc/2. The correlators 15A and 15B correlate sampled signals which were sampled during a period equivalent to one cycle of the PN signal with the reference PN signals. Since one cycle of the PN signal is N, the number of each of the sampled signals 14a and 14b having been subjected to sampling during a period equivalent to one cycle of the PN signal is 2N. Sampling per twice chip results in a sampled signal having the same PN signal adjacent to the sampled signals 14a and 14b, and the correlators 15A and 15B generate correlation signals as expressed below, making use of 2N sampled signals having been inputted thereto. ##EQU1## where .gamma.i (i=1, . . . , 2N) is 2N sampled signals to be inputted, cj is the reference PN signal having a value of either -1 or +1, and [n] is the maximum integer which does not exceed n. The correlators can be said to be circuits which generate an auto correlation of the PN signal.
Next, referring to FIG. 2, the relationship between the base band signals 13a and 13b subjected to sampling performed twice per chip and a sampling timing will be described. Presume that a time t=iTC (i=1, 2, . . . ) is the center of each chip waveform, that times when two sampling timings of each chip respectively offset +.DELTA.t from two timings (iTc-Tc/4) and (iTc+Tc/4) symmetrical with the center of the each chip waveform are defined as t.sub.1 (=iTc-Tc/4+.DELTA.t) and t.sub.2 (=iTc+Tc/4+.DELTA.t), and that the sampling circuits 14A and 14B sample the analog quantity of input signals as they are, the sampled signals 14a and 14b sampled at the timing t.sub.2 are respectively expressed as follows. EQU d(t.sub.2)c(t.sub.2)h(Tc/4+.DELTA.t)cos.theta. (4) EQU d(t.sub.2)c(t.sub.2)h(Tc/4+.DELTA.t)sin.theta. (5)
where h (t.sub.0) represents a unit pulse waveform of a chip which is characterized by the characteristic of the low pass filters 13A and 13B, and h(.sub.0) corresponds to the center of the chip waveform.
If c(t.sub.2) is rewritten into ci (having a value of -1 or +1), the sampled signals 14a and 14b sampled at the timing of t.sub.2 can be expressed as EQU d(t.sub.2)c.sub.i h(Tc/4+.DELTA.t)cos.theta. (6) EQU d t.sub.2)c.sub.i h(Tc/4+.DELTA.t)sin.theta. (7)
The relationship between the correlation signals 15a and 15b and the sampling timing will now be explained with reference to FIG. 3.
At a sampling timing ts (=kNTc+Tc/4+.DELTA.t), (k=1, 2, . . . ), the correlators 15A and 15B correlate 2N sampled signals 14a and 14b from a timing tu (={(k-1) N+1} Tc-Tc/4+.DELTA.t) to a timing ts (=kNTc+Tc/4+.DELTA.t).
Each of 2N number of sampled signals 14a and 14b contains two PN signal components from c.sub.1 to c.sub.N. Since each sampled signal contains c.sub.1, c.sub.1, c.sub.2, c.sub.2, . . . , c.sub.N, c.sub.N, the PN signal component comprised in the received signal 1 at the sampling time ts is matched with the reference PN signal.
Accordingly, the correlation signals 15a and 15b will be signals from which the PN signal components have been respectively eliminated, as expressed by c.sub.i .times.c.sub.1 =1. ##EQU2##
If one information data bit is spread by the PN signal, the information data does not change within one cycle of the PN signal. Hence, if d ((k-1) NTc+iTc-Tc/4+.DELTA.t) and d((k-1) NTc+iTc+Tc/4+66 t) are represented as d.sub.k (having a value of -1 or +1), the correlation signals 15a and 15b at the sampling timing ts when the PN signal components included in the received signal 1 and the reference PN signal are completely matched with each other can be expressed as EQU Nd.sub.k {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}cos.theta. (10) EQU Nd.sub.k {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}sin.theta. (11)
Contrary to this, the correlation signals 15a and 15b at a timing when the PN signal components included in the received signal 1 are out of phase with the reference PN signal becomes -1 when the PN signal is M-sequence. Hence, respectively, respective these correlation signals can be represented as EQU -d.sub.k {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}cos.theta. (12) EQU -d.sub.k {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}sin.theta. (13)
Since d.sub.k .times.d.sub.k -1, the information data component d.sub.k is eliminated from the correlation signals 15a and 15b by squaring the signals by means of the square generators 16A and 16B. Thereafter, the correlation pulse signal 24 is produced by adding the correlation signals 15a and 15b at the adder 23.
As a result of this, the correlation pulse signal 24 does not include any information data components, and is represented by the following equation at the timing ts when the PN signal components included in the received signal 1 is completely matched with the reference PN signal. EQU N.sup.2 {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}.sup.2 ( 14)
Contrary, the correlation pulse signal 24 is represented by the following equation at a timing when the PN signal components included in the received signal 1 is out of phase with the reference PN signal. EQU {h(-Tc/4+.DELTA.t)+h(+Tc/4+.DELTA.t)}.sup.2 ( 15)
Since a PN signal having a large cycle of N is commonly used in spread spectrum communications, there consequently arises a large difference between the amplitude of the correlation pulse signal 24 produced at the time when the received PN signal is in phase with the reference PN signal and the amplitude of the correlation pulse signal 24 produced at the time when these signals are out of phase with each other.
The decision device 25 decides a generation timing of the acquired signal 3 making use of the above mentioned characteristics, and outputs the acquired signal 3, thereby completing the initial acquisition operation.
Subsequently, the acquired signal 3 thus generated by the initial acquisition circuit is transmitted to the tracking circuit 4 shown in FIG. 16, and thereafter tracking of the PN signal included in the received signal 1 is effected. A conventional tracking circuit 4 will be described hereinbelow.
FIG. 18 shows the structure of the conventional tracking circuit used in the spread spectrum demodulator shown in the aforementioned documents written by M. K. Simon.
In FIG. 18, reference numerals 26A and 26B designate multipliers for multiplying the received signal 1 by an early PN reference signal 39 and a late PN reference signal 40 respectively, both being generated by a PN signal generator 38; 27A and 27B, band-pass filters (BPF) having a center frequency equivalent to the carrier frequency of the received signal 1; 28A and 28B, square generators; 34, a subtracter for producing a difference of levels between the output of the square generators 28A and 28B; 35 and 35a, error signals; 36, a loop filter; 37, a voltage controlled oscillator for generating a reproduced clock 37a by varying an output frequency by use of a voltage of the error signal 35a; 38, a PN signal generator for generating the early and the late PN reference signals 39, 40 from the reproduced clock 37a; and 41, a delay element for causing the Tc/2 period delay of the early PN reference signal 39.
A description will be given of the tracking circuit shown in FIG. 18.
Upon receipt of the acquired signal 3, the PN signal generator 38 starts outputting both the early PN reference signal 39 and the late PN reference signal 40. The early PN reference signal 39 is advanced in phase by Tc/2 with respect to the punctual PN reference signal 5, whilst the late PN reference signal 40 is delayed in phase by a Tc/2 with respect to the punctual PN reference signal 5.
Assume that the punctual PN reference signal 5 is delayed in phase by a time, .tau., with respect to the phase of a PN signal c(t) included in the received signal 1. The punctual PN reference signal 5 designated by S.sub.R (t), the early PN reference signal 39 designated by S.sub.E (t), and the late PN reference signal 40 by S.sub.L (t) are characterized by the following equations. EQU S.sub.R (t)=c(t-.tau.) (16) EQU S.sub.E (t)=S.sub.R (t+Tc/2)=c(t-.tau.+Tc/2) (17) EQU S.sub.L (t)=S.sub.R (t-Tc/2)=c(t-.tau.-Tc/2) (18)
The received signal 1 (=d (t) c (t) cos.omega..sub.0 t) is divided into two, and one half is multiplied by the early PN reference signal S.sub.E (t) by means of the mixer 26A, and the thus obtained signal is inputted in the band-pass filter 27A. The band-pass filter 27A passes a signal whose spectrum band width is within that of the information data d(t). This filter is inserted in order to eliminate unwanted noise components received together with the received signal 1. Instantaneous value components c (t) c (t-.tau.+Tc/2) included in the signal outputted from the band-pass filter 27A are eliminated by the characteristic of a loop which constitutes the tracking circuit that suppresses a high frequency component, leaving an average of a result from the multiplication of the PN signal contained in the signal outputted from the band pass filter 27A. This average value results in a signal expressed as below. EQU d(t)c(t)c(t-.tau.+Tc/2 cos.omega..sub.0 t (19)
where c (t) c (t-.tau.+Tc/2) represents a time average of c (t) c (t-.tau.+Tc/2 component.
The time average of the c (t) c (t-.tau.+Tc/2) represents the autocorrelation function of the PN signal, and shows a maximum value rate Tc/2 and a minimum value -1 at .vertline..tau.-Tc/2 .vertline..gtoreq.Tc.
Assume that the autocorrelation function is R.sub.E (.tau.), then the R.sub.E (.tau.) is expressed as ##EQU3##
Squaring the output signal from the band-pass filter 27A by means of the square generator 28A results in elimination of the information data component d(t). Despite the fact that squaring the carrier frequency component w.sub.0 causes generation of a DC component and a 2w.sub.0 component, the characteristic of the loop which constitutes the tracking circuit for suppressing the high frequency component allows the elimination of the 2w.sub.0 components, eventually the DC component simply appearing at the output of the band-pass filter 27A.
As a result of this, the signal 29a outputted from the square loop 28A becomes {R.sub.E (.tau.)}.sup.2. The other half of the divided received signal 1 is multiplied by the late PN reference signal S.sub.L (t) by means of the mixer 26B, and then transmitted to the band-pass filter 27B. The output signal of the bandpass filter 27B is then expressed as EQU d(t)c(t)c(t-.tau.Tc/2)cos.omega..sub.0 t(k=1, 2, . . . ) (21)
Suppose the autocorrelation function of the PN signal which is included in the above signal is R.sub.L (.tau.), the R.sub.L (.tau.) will be represented as ##EQU4##
Squaring the output signal from the band-pass filter 27B by means of the square loop 28B results in the signal 29b represented as {R.sub.L (.tau.)}2.
The subtracter 34 generates the error signal 35 from the signals 29a and 29b, and this error signal is then subjected to elimination of unwanted noise by the loop filter 36, producing the error signal 35a. The error signals 35 and 35a are expressed by the following equation. EQU D(.tau.)={R.sub.E (.tau.)}.sup.2 -{R.sub.L (.tau.)}.sup.2 ( 23)
FIG. 19 shows the amplitude characteristic of the error signals 35 and 35a. Thus, the error signals are configured in S shape symmetry with respect to the origin of coordinates, and have a characteristic to produce D (.tau.)&gt;0 with .tau.&gt;0 and D (.tau.)&lt;0 with .tau.&lt;0. Such characteristics are hereinafter referred to as a time discrimination characteristic.
The voltage controlled oscillator 37 produces an output of the reproduced clock 37a having a frequency corresponding to the amplitude of the error signals. In response to the output from the voltage controlled oscillator 37, the PN signal generator 38 outputs both the early and late PN reference signal 39 and 40.
This operation is effected in such a manner as to maintain the value .tau. of 0, and hence the early PN reference signals 39 is in synchronism with the PN signal included in the received signal 1 which is phase advanced by Tc/2, whereas the late PN reference signal 40 is in synchronism in phase delayed by Tc/2.
The early PN reference signal 39 is delayed by a period of Tc/2 by means of the delay element 41 and outputted as the punctual PN reference signal 5 completely in phase with (.tau.=0) the PN signal components included in the received signal 1.
As mentioned above, the PN signal components included in the received signal 1 shown in FIG. 16 are eliminated (despread) using the punctual PN reference signal 5 which was generated by the tracking circuit, and subsequently the information data is demodulated by means of the conventional PSK demodulator 9.
In the conventional spread spectrum demodulator apparatus having the foregoing structure, the initial acquisition circuit performs initial acquisition using a received signal from which PN signal components have been eliminated, and the tracking circuit exercises tracking using the received signal from which the PN signal components have been eliminated in the same manner.
Thus, in the conventional spread spectrum demodulator, the operation of eliminating a PN signal component included in the received signal which is performed by the initial acquisition circuit and the tracking circuit independently of each other has resulted in a bulky apparatus with a complex structure.
In the case where the conventional spread spectrum demodulator finds application in the fields of mobile satellite communications such as mobile telephones and the like, a plurality of signals (hereinafter referred to as a multipath waveform) having different propagation delays are received at the same time, and the level of each received signal varies with time. If tracking is effected for only one of these signals, a sharp drop in the level of the received signal of interest will occur, even down to 0, when the received signal that is the target of the tracking is interfered with obstacles such as buildings. Eventually, a lack of error signals which are necessary in tracking, namely a drop of the amplitude of the error signals to zero, causes asynchronization (what is called locking-off) due to the improper tracking. The locking-off makes demodulation of information data impossible.